Shift register and image display apparatus containing the same

ABSTRACT

A shift register has an output stage formed by a first transistor connected between an output terminal and a first clock terminal and a second transistor connected between the output terminal and a ground. Third and fourth transistors are connected in series between the gate of the first transistor (first node) and the ground. A second node between the third and fourth transistors is connected to a power source via a fifth transistor. The fifth transistor has its gate connected to the first node. Accordingly, when the third and fourth transistors are turned off to raise the first node in level, the fifth transistor is turned on, whereby a predetermined voltage is applied to the second node.

CROSS REFERENCE TO RELATED APPLICATION

This application is a divisional application of application Ser. No.11/258,058, filed Oct. 26, 2005, now U.S. Pat. No. 7,289,593.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a shift register, and moreparticularly, to a shift register for use as a scanning-line drivingcircuit for an image display apparatus or the like, which is formed byfield effect transistors of the same conductivity type only.

2. Description of the Background Art

An image display apparatus (hereinafter referred to as a “displayapparatus”) such as a liquid crystal display includes a display panel inwhich a plurality of pixels are arrayed in a matrix. A gate line(scanning line) is provided for each row of pixels (pixel line), andgate lines are sequentially selected and driven in a cycle of onehorizontal period of a display signal, so that a displayed image isupdated. As a gate-line driving circuit (scanning-line driving circuit)for sequentially selecting and driving pixel lines, i.e., gate lines, ashift register for performing a shift operation in one frame period of adisplay signal can be used.

To reduce the number of steps in the manufacturing process of a displayapparatus, such shift register used as the gate-line driving circuit ispreferably formed by field effect transistors of the same conductivitytype only. Accordingly, various types of shift registers formed by N- orP-type field effect transistors only and display apparatuses containingsuch shift registers have been proposed (e.g., U.S. Pat. No. 5,222,082,and Japanese Patent Application Laid-Open Nos. 2002-313093, 2002-197885and 2004-103226). As a field effect transistor, a metal oxidesemiconductor (MOS) transistor, a thin film transistor (TFT), or thelike is used.

A conventional shift register has a problem caused by a leakage currentat a node (specifically, nodes P1 and P2 shown in FIG. 2 of theabove-mentioned U.S. Pat. No. 5,222,082) to which a gate electrode of atransistor of an output stage is connected.

For instance, when a leakage current occurs at a gate electrode node(P1) of a transistor connected between an output terminal of a shiftregister and a clock terminal defining an output signal output from theoutput terminal, an impedance of the transistor when discharge takesplace at the output terminal increases, and the time required for thedischarge thus increases. Therefore, the fall time of the output signalincreases, causing the output signal to be unable to follow a clocksignal input to the clock terminal. As a result, an increased fall timeof the output signal in the gate-line driving circuit of a displayapparatus causes a problem in that a plurality of gate lines are drivenat the same time, resulting in failure to achieve a normal display(which will be described later in detail).

Further, when a leakage current occurs at a gate electrode node (P2) ofa transistor connected between the output terminal and a referencevoltage terminal of the shift register, an impedance of the transistorin the on state (conducting state) increases. That is, an outputimpedance of the shift register increases, arising a concern that thepotential at the output terminal may become unstable. In the case wherethe output signal from the gate-line driving circuit thus becomesunstable, the problem of failure to achieve a normal display also arises(which will be described later in detail).

A shift register described in the above-mentioned JP2002-313093 includesinversion-preventing circuits (transistors T7, T8) connected to the gateelectrode node (n2) of an NMOS transistor (transistor T2) connectedbetween an output terminal and a power source for stabilizing thepotential at the node. The NMOS transistor needs to be held off (cutoffstate) in a period during which an output line is in low level. Theinversion-preventing circuits prevent the NMOS transistor fromunnecessarily turning on due to level transition of the output lineduring the period in which the output line is in the low level, whichsolves another problem different from the above-described one.

SUMMARY OF THE INVENTION

An object of the present invention is to provide a shift registercapable of preventing a malfunction caused by a leakage current and animage display apparatus containing such shift register.

According to a first aspect of the invention, the shift registerincludes an input terminal and an output terminal, first and secondclock terminals, first, second and third voltage terminals, a firsttransistor, a second transistor, a first node, a second node and adriving section. First and second clock signals are input to the firstand second clock terminals, respectively. The first and second clocksignals are shifted in phase from each other. First, second and thirdvoltages are applied to the first, second and third voltage terminals,respectively. The first transistor is connected between the outputterminal and the first clock terminal. The second transistor isconnected between the output terminal and the first voltage terminal. Acontrol electrode of the first transistor is connected to the firstnode. A control electrode of the second transistor is connected to thesecond node. The driving section applies the first voltage to the firstnode and a voltage corresponding to the third voltage to the second nodein synchronization with the second clock signal, while applying avoltage corresponding to the second voltage to the first node and thefirst voltage to the second node in response to an input signal input tothe input terminal. The driving section includes a third transistor forapplying the first voltage to the first node, the third transistorhaving one main electrode connected to the first node and a controlelectrode connected to the second node. The driving section isconfigured such that a predetermined voltage other than the firstvoltage is applied to a third node which is the other main electrode ofthe third transistor in a period during which the third transistor is ina cutoff state.

It is possible to control a leakage current at the first node of theshift register, which prevents the first node from dropping in voltagelevel while being charged. As a result, an output signal in an activestate of the output terminal is ensured to follow the first clocksignal, which achieves an improved operational reliability. Therefore, adisplay apparatus containing the shift register as a scanning-linedriving circuit can achieve a normal display while preventing amalfunction.

According to a second aspect of the invention, the shift registerincludes an input terminal and an output terminal, first and secondclock terminals, first, second and third voltage terminals, a firsttransistor, a second transistor, a first node, a second node and adriving section. First and second clock signals are input to the firstand second clock terminals, respectively. The first and second clocksignals are shifted in phase from each other. First, second and thirdvoltages are applied to the first, second and third voltage terminals,respectively. The first transistor is connected between the outputterminal and the first clock terminal. The second transistor isconnected between the output terminal and the first voltage terminal. Acontrol electrode of the first transistor is connected to the firstnode. A control electrode of the second transistor is connected to thesecond node. The driving section applies the first voltage to the firstnode and a voltage corresponding to the third voltage to the second nodein synchronization with the second clock signal, while applying avoltage corresponding to the second voltage to the first node and thefirst voltage to the second node in response to an input signal input tothe input terminal. The driving section includes third and fourthtransistors connected in series between the second node and the firstvoltage terminal, each of the third and fourth transistors having acontrol electrode connected to the input terminal. The driving sectionis configured such that a predetermined voltage other than the firstvoltage is applied to a third node which is a connection node betweenthe third and fourth transistors in a period during which the third andfourth transistors are in a cutoff state.

It is possible to control a leakage current at the second node, whichprevents the second node from dropping in voltage level while beingcharged. Accordingly, an impedance of the second transistor in aninactive state of the output terminal, that is, an output impedance ofthe shift register is prevented from increasing, which achieves animproved operational reliability. Therefore, a display apparatuscontaining the shift register as a scanning-line driving circuit canachieve a normal display while preventing a malfunction.

According to a third aspect of the invention, the shift registerincludes an input terminal and an output terminal, first and secondclock terminals, first, second and third voltage terminals, a firsttransistor, a second transistor, a first node, a second node and adriving section. First and second clock signals are input to the firstand second clock terminals, respectively. The first and second clocksignals are shifted in phase from each other. First, second and thirdvoltages are applied to the first, second and third voltage terminals,respectively. The first transistor is connected between the outputterminal and the first clock terminal. The second transistor isconnected between the output terminal and the first voltage terminal. Acontrol electrode of the first transistor is connected to the firstnode. A control electrode of the second transistor is connected to thesecond node. The driving section applies the first voltage to the firstnode and a voltage corresponding to the third voltage to the second nodein synchronization with the second clock signal, while applying avoltage corresponding to the second voltage to the first node and thefirst voltage to the second node in response to an input signal input tothe input terminal. The driving section includes a compensation circuitfor charging the second node to a level at which a conducting state ofthe second transistor is kept in synchronization with the first clocksignal during a period in which the second transistor is in theconducting state.

A leakage current, if occurred at the second node having been charged,is compensated for in an inactive state of the output terminal, whichbrings the second transistor into the conducting state. Accordingly, animpedance of the second transistor in an inactive state of the outputterminal, that is, an output impedance of the shift register isprevented from increasing, which achieves an improved operationalreliability. Therefore, a display apparatus containing the shiftregister as a scanning-line driving circuit can achieve a normal displaywhile preventing a malfunction.

These and other objects, features, aspects and advantages of the presentinvention will become more apparent from the following detaileddescription of the present invention when taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic block diagram illustrating the configuration of adisplay apparatus according to the present invention;

FIG. 2 is a block diagram illustrating the configuration of a gate-linedriving circuit of the display apparatus according to a first preferredembodiment of the invention;

FIG. 3 is a circuit diagram illustrating the configuration of a unitshift register according to the first preferred embodiment;

FIG. 4 is a timing chart showing an operation of the unit shift registeraccording to the first preferred embodiment;

FIG. 5 is a timing chart showing an operation of the gate-line drivingcircuit of the display apparatus according to the first preferredembodiment;

FIG. 6 is a graph showing the effects achieved by the first preferredembodiment;

FIGS. 7 and 8 are diagrams each illustrating an example of power circuitconnected to a third power terminal according to the first preferredembodiment;

FIG. 9 is a circuit diagram illustrating the configuration of a unitshift register according to a second preferred embodiment of theinvention;

FIG. 10 is a circuit diagram illustrating the configuration of a unitshift register according to a third preferred embodiment of theinvention;

FIG. 11 is a circuit diagram illustrating the configuration of a unitshift register according to a fourth preferred embodiment of theinvention;

FIG. 12 is a circuit diagram illustrating the configuration of a unitshift register according to a fifth preferred embodiment of theinvention,

FIG. 13 is a block diagram illustrating the configuration of a gate-linedriving circuit of a display apparatus according to a sixth preferredembodiment of the invention;

FIG. 14 is a timing chart showing an operation of the gate-line drivingcircuit of the display apparatus according to the sixth preferredembodiment;

FIG. 15 is a circuit diagram illustrating the configuration of a unitshift register according to the sixth preferred embodiment;

FIG. 16 is a circuit diagram illustrating the configuration of a unitshift register according to a seventh preferred embodiment of theinvention;

FIG. 17 is a circuit diagram illustrating the configuration of a unitshift register according to an eighth preferred embodiment of theinvention;

FIGS. 18 and 19 are diagrams each illustrating an example of powercircuit connected to a seventh power terminal according to the eighthpreferred embodiment;

FIG. 20 is a circuit diagram illustrating the configuration of a unitshift register according to a ninth preferred embodiment of theinvention;

FIG. 21 is a circuit diagram illustrating the configuration of a unitshift register according to a tenth preferred embodiment of theinvention;

FIG. 22 is a timing chart showing an operation of the unit shiftregister according to the tenth preferred embodiment;

FIG. 23 is a circuit diagram illustrating the configuration of a unitshift register according to a variant of the tenth preferred embodiment;

FIG. 24 is a circuit diagram illustrating the configuration of a unitshift register according to an eleventh preferred embodiment of theinvention;

FIG. 25 is a circuit diagram illustrating the configuration of a unitshift register according to a twelfth preferred embodiment of theinvention;

FIG. 26 is a circuit diagram illustrating the configuration of a unitshift register according to a thirteenth preferred embodiment of theinvention;

FIG. 27 is a circuit diagram illustrating the configuration of a unitshift register according to a fourteenth preferred embodiment of theinvention;

FIG. 28 is a circuit diagram illustrating the configuration of a unitshift register according to a fifteenth preferred embodiment of theinvention;

FIG. 29 is a circuit diagram illustrating the configuration of a unitshift register according to a sixteenth preferred embodiment of theinvention;

FIG. 30 is a circuit diagram illustrating the configuration of a unitshift register according to a seventeenth preferred embodiment of theinvention; and

FIG. 31 is a circuit diagram illustrating the configuration of a unitshift register according to an eighteenth preferred embodiment of theinvention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Preferred embodiments of the present invention will be describedhereinbelow referring to the accompanied drawings. To avoid repeated andredundant description, elements having the same or correspondingfunctions are indicated by the same reference characters in thedrawings.

First Preferred Embodiment

FIG. 1 is a schematic block diagram illustrating the configuration of adisplay apparatus according to the present invention. The overallconfiguration of a liquid crystal display 10 is shown as anrepresentative example of the display apparatus.

The liquid crystal display 10 is provided with a liquid crystal arraypart 20, a gate-line driving circuit (scanning-line driving circuit) 30and a source driver 40. As will be described later explicitly, a shiftregister according to the present embodiment is mounted on the gate-linedriving circuit 30.

The liquid crystal array part 20 includes a plurality of pixels 25arrayed in a matrix. The columns of pixels (hereinafter also referred toas “pixel lines”) are respectively provided with gate lines GL1, GL2, .. . (hereinafter also generically referred to as a “gate line GL”), andthe rows of pixels (hereinafter also referred to as “pixel rows”) arerespectively provided with data lines DL1, DL2, . . . (hereinaftergenerically referred to as a “data line DL”). FIG. 1 representativelyshows pixels 25 of the first and second rows in the first column andcorresponding gate line GL1 and data lines DL1 and DL2.

Each pixel 25 has a pixel switching device 26 disposed between acorresponding data line DL and a pixel node Np, and a capacitor 27 and aliquid crystal display device 28 connected in parallel between the pixelnode Np and a common electrode node NC. The crystal orientation in theliquid crystal display device 28 changes depending on the potentialdifference between the pixel node Np and common electrode node NC, andin response to this change, the display luminance of the liquid crystaldisplay device 28 changes. Accordingly, the luminance of each pixel 25can be controlled by a display voltage transmitted to the pixel node Npvia the data line DL and pixel switching device 26. In other words, anintermediate potential difference between a potential differencecorresponding to the maximum luminance and a potential differencecorresponding to the minimum luminance is applied between the pixel nodeNp and common electrode node NC, whereby halftone luminance can beobtained. Therefore, setting display voltages stepwise, grayscaleluminance can be obtained.

The gate-line driving circuit 30 sequentially selects and drives a gateline GL in a predetermined scanning cycle. Each pixel switching device26 has its gate electrode connected to a corresponding gate line GL.While a certain gate line GL is selected, the pixel switching device 26is brought into the conducting state in each pixel 25 connected to theselected gate line GL, whereby the pixel node Np is connected to acorresponding data line DL. Then, the display voltage transmitted to thepixel node Np is held by the capacitor 27. Generally, the pixelswitching device 26 is constructed from a TFT formed on a substrate ofthe same insulator as the liquid crystal display device 28 (a glasssubstrate, a resin substrate or the like).

The source driver 40 is provided to output display voltages set stepwiseby a display signal SIG which is an N-bit digital signal, to the dataline DL. As an example, the display signal SIG is assumed to be a 6-bitsignal including display signal bits D0 to D5. With such 6-bit displaysignal SIG, 2⁶=64 levels of gray can be displayed in each pixel 25.Further, a display of approximately 260 thousand colors can be achievedby forming one color display unit by three pixels of R (Red), G (Green)and B (Blue).

As shown in FIG. 1, the source driver 40 includes a shift register 50,data latch circuits 52, 54, a gradation voltage generating circuit 60, adecoder circuit 70 and an analog amplifier 80.

In the display signal SIG, the display signal bits D0 to D5corresponding to the display luminance of respective pixels 25 areserially generated. In other words, each of the display signal bits D0to D5 at each timing indicates the display luminance of any one pixel 25in the liquid crystal array part 20.

The shift register 50 gives an instruction to the data latch circuit 52to capture the display signal bits D0 to D5 in synchronization with acycle during which the settings of the display signal SIG are changed.The data latch circuit 52 sequentially captures serially-generateddisplay signals SIG to latch display signals SIG for one pixel line.

A latch signal LT input to the data latch circuit 54 is activated withtiming when display signals SIG for one pixel line are captured by thedata latch circuit 52. In response to this, the data latch circuit 54captures the display signals SIG for one pixel line latched by the datalatch circuit 52 at that time.

The gradation voltage generating circuit 60 is formed by 63 resistordividers connected in series between a high voltage VDH and a lowvoltage VDL, for generating 64 levels of gradation voltages V1 to V64,respectively.

The decoder circuit 70 decodes display signals SIG latched by the datalatch circuit 54, and based on the result of decoding, selects voltagesto be respectively output to decoder output nodes Nd1, Nd2, . . .(generically referred to as a “decoder output node Nd”) from among thegradation voltages V1 to V64, and outputs the selected voltages.

As a result, display voltages (selected from among the gradationvoltages V1 to V64) corresponding to the display signals SIG for onepixel line latched by the data latch circuit 54 are output to thedecoder output node Nd at the same time (in parallel). FIG. 1representatively shows the decoder output nodes Nd1 and Nd2corresponding to the data line DL1 of the first row and the data lineDL2 of the second row, respectively.

The analog amplifier 80 outputs analog voltages corresponding to displayvoltages output from the decoder circuit 70 to the decoder output nodesNd1, Nd2 . . . , to the data lines DL1, DL2, . . . , respectively.

The source driver 40 repeatedly outputs display voltages correspondingto a series of display signals SIG for one pixel line to the data lineDL in a predetermined scanning cycle, and the gate-line driving circuit30 sequentially drives the gate lines GL1, GL2, . . . in synchronizationwith the scanning cycle. Accordingly, image display based on the displaysignals SIG is provided on the liquid crystal array part 20.

FIG. 1 shows an example of the liquid crystal display 10 with thegate-line driving circuit 30 and source driver 40 formed integrally withthe liquid crystal array part 20, however, the gate-line driving circuit30 and source driver 40 may be provided as an external circuit of theliquid crystal array part 20.

The configuration of the gate-line driving circuit 30 according to thepresent invention will now be described in detail. FIG. 2 shows theconfiguration of the gate-line driving circuit 30 according to thepresent embodiment. The gate-line driving circuit 30 includes aplurality of shift registers SR1, SR2, SR3, SR4, . . . connected incascade (for ease of description, each of the cascaded shift registersSR1, SR2, . . . will generically be called a “unit shift register SR”).Each unit shift resister SR is provided for one pixel line, i.e., eachgate line GL.

A clock generator 31 shown in FIG. 2 is provided to input three phaseclock signals C1, C2 and C3, shifted in phase with each other, to theunit shift register SR of the gate-line driving circuit 30. The clocksignals C1, C2 and C3 are controlled to be sequentially activated insynchronization with the scanning cycle of the display apparatus.

Each unit shift register SR includes an input terminal IN, an outputterminal OUT, and first and second clock terminals A and B. As shown inFIG. 2, two of the clock signals C1, C2 and C3 output from the clockgenerator 31 are supplied to the clock terminals A and B of each unitshift register SR, respectively. An input signal called a start pulse isinput to the input terminal IN of the unit shift register SR1 of thefirst stage. Input to the input terminal IN of each of the unit shiftregisters SR of the second and following stages is an output signaloutput to the output terminal OUT of the immediately preceding stage.The output signal of each unit shift register SR is output to the gateline GL as a horizontal (or vertical) scanning pulse.

With the gate-line driving circuit 30 of such configuration, each unitshift register SR outputs an input signal received from the immediatelypreceding stage (output signal of the immediately preceding stage) to acorresponding gate line GL and a unit shift register SR of theimmediately succeeding stage while shifting the input signal insynchronization with the clock signals C1, C2 and C3 (the operation ofthe unit shift register SR will be described later in detail). As aresult, a series of unit shift registers SR serve as a so-calledgate-line driving unit for sequentially activating gate lines GL withtiming based on the predetermined scanning cycle.

FIG. 3 is a circuit diagram illustrating the configuration of the unitshift register SR according to the present embodiment. As the respectiveunit shift registers SR have substantially the same configuration, theconfiguration of one unit shift register SR will be representativelydescribed hereinbelow. Transistors constituting the unit shift registerSR shall be field effect transistors of the same conductivity type, andin the present embodiment, all the transistors shall be N-type TFTs.

The unit shift register SR includes the input terminal IN, outputterminal OUT, a first clock terminal A, second clock terminal B, areference voltage terminal serving as a first voltage terminal to whicha first voltage is applied, a first power terminal s1 serving as asecond voltage terminal to which a predetermined second voltage isapplied and a second power terminal s2 serving as a third voltageterminal to which a predetermined third voltage is applied. The presentembodiment illustrates a case in FIG. 3 in which, assuming the voltageon the shift register side as a reference for ease of description, thereference voltage terminal is connected to a ground GND (0V) and thefirst and second power terminals s1 and s2 are both connected to a powersource VDD (that is, the first voltage is 0V, and the second and thirdvoltages are the source voltage VDD in this embodiment). In practicaluse, the voltage on the pixel side is assumed as a reference, and in theexample of FIG. 3, a voltage of 17V is applied to the first and secondpower terminals s1 and s2 and a voltage of −12V is applied to thereference voltage terminal (that is, in this practical example, thefirst voltage is −12V, and the second and third voltages are the sourcevoltage of 17V). In other words, the present embodiment will describethat the unit shift register SR operates such that a corresponding gateline GL is at the source voltage VDD when selected and 0V when notselected, however, in practical use, the unit shift register SR operatessuch that a corresponding gate line GL is in a positive level (e.g.,17V) when selected and negative level (e.g., −12V) when not selected.

The output stage of the unit shift register SR is formed by a transistorT1 (first transistor) connected between the output terminal OUT andfirst clock terminal A and a transistor T2 (second transistor) connectedbetween the output terminal OUT and ground GND (reference voltageterminal). As shown in FIG. 3, the transistor T1 has its gate (controlelectrode) connected to a node N1 (first node), and the transistor T2has its gate connected to a node N2 (second node). A transistor T3 isconnected between the node N1 and first power terminal s1 (power sourceVDD), and transistors T4 and T7 are connected in series between the nodeN1 and ground GND.

The transistors T4 and T7 are provided to apply a reference voltage (theground voltage GND) to the node N1. As shown in FIG. 3, the transistorT4 has its drain which is one main electrode connected to the node N1and its source which is the other main electrode connected to thetransistor T7. The transistor T7 is connected between a node N3 andground GND. The transistors T4 and T7 have their gates connected incommon to the node N2. A node for the source of the transistor T4 (here,a connection node between the transistors T4 and T7) is defined as thenode N3.

The unit shift register SR further includes a third power terminal s3,and a transistor T8 is connected between the third power terminal s3 andnode N3. In the present embodiment, a predetermined power source VDM isconnected to the third power terminal s3, and the transistor T8 has itsgate connected to the node N1. That is, the transistor T8 is turned onwhen the node N1 rises in level to apply the voltage at the third powerterminal s3 (output voltage from power source VDM) to the node N3.

A transistor T5 is connected between the node N2 and second powerterminal s2 (power source VDD), and a transistor T6 is connected betweenthe node N2 and ground GND (reference voltage terminal). The inputterminal IN is connected to the gates of the transistors T3 and T6, andthe second clock terminal B is connected to the gate of the transistorT5. The unit shift register SR according to the present embodiment isconfigured as described above.

As will be apparent from the above-mentioned U.S. Pat. No. 5,222,082 andJP2002-313093, a conventional unit shift register has only onetransistor connected between the node N1 and reference voltage terminal(ground GND) (cf. transistor 19 shown in FIG. 2 of U.S. Pat. No.5,222,082). In other words, the conventional unit shift register isconfigured such that the transistor T4 has its source (node N3) directlyconnected to the ground GND removing the transistors T7 and T8 from thecircuit shown in FIG. 3 of the present application.

In the present embodiment, the transistors T3 to T8 constitute a drivingsection for driving the unit shift register SR. This driving sectionapplies the voltage at the reference voltage terminal (ground GND) tothe node N1 and the voltage at the second power terminal s2 (powersource VDD) to the node N2 in synchronization with a clock signal inputto the second clock terminal B, while applying the voltage at the firstpower terminal s1 (power source VDD) to the node N1 and the voltage atthe reference voltage terminal (ground GND) to the node N2 in responseto an input signal input to the input terminal IN. The operation of theunit shift register SR according to the present embodiment includingthis driving section will now be described specifically.

FIG. 4 is a timing chart showing a normal operation of the unit shiftregister SR according to the present embodiment. Here, it is assumedthat the clock signal C1 is input to the first clock terminal A of theunit shift register SR and the clock signal C3 is input to the secondclock terminal B (this case corresponds to, e.g., the unit shiftregisters SR1 and SR4 shown in FIG. 2). The voltage level (hereinafterbriefly referred to as a “level”) output from the power sources VDD andVDM will hereinafter be indicated as “VDD” (VDD>0) and “VDM” (VDM>0),respectively.

As shown in FIG. 4, when the clock signal C3 (at the second clockterminal B) transitions from 0V to VDD at time t₀, then, the transistorT5 is turned on (conducting state). At this time, the input terminal INis at 0V and thus, the transistor T6 is held off (cutoff state). Thus,the node N2 is charged to VDD−Vth (Vth: threshold voltage oftransistor). Then, the transistors T4 and T7 are both turned on, and thenodes N1 and N3 transition to 0V. Following the transition of the nodeN1 to 0V, the transistors T1 and T8 are turned off.

As described, when the driving section applies the source voltage VDD tothe node N1 and the ground voltage GND to the node N2, the transistor T1is turned off and the transistor T2 is turned on, which brings the gateline GL into an inactive state (non-selective state) having a lowimpedance.

Next, when the clock signal C3 returns to 0V at time t₁, the transistorT5 is turned off, while the transistor T6 is held off. Thus, the node N2is kept at VDD−Vth.

At time t₂ when an input signal is input to the input terminal IN, theinput terminal IN transitions to VDD, and the transistors T3 and T6 arethen turned on. Accordingly, the node N2 is discharged to transition to0V, and the transistors T2, T4 and T7 are turned off. As the transistorT3 is on, the node N1 is charged this time to VDD−Vth. Accordingly, thetransistor T1 is turned on (since VDD>>Vth generally holds, VDD−Vth>Vthholds).

As described, when the driving section applies the ground voltage GND tothe node N1 and the source voltage VDD to the node N2, the transistor T1is turned on and the transistor T2 is turned off. At this time, theoutput terminal OUT does not transition from 0V since the clock signalC1 (at the first clock terminal A) is 0V. That is, the gate line GLremains in the inactive state having a low impedance at this point oftime.

In the present embodiment, the transistor T8 is also turned on at thistime, and the source voltage VDM is applied to the node N3. The sourcevoltage VDM is assumed to be at such a level that the transistor T8operates in a non-saturation region. In this case, the node N3transitions to VDM (when VDM is at such a level that the transistor T8operates in a saturation region, the node N3 transitions to VDD−2×Vth).

When the input terminal IN returns to 0V at time t₃, the transistors T3and T6 are turned off, however, the transistors T4 and T7 are held off,causing the node N1 to be kept at VDD−Vth (where a leakage current isdisregarded).

Next, at time t₄, the clock signal C1 input to the first clock terminalA transitions from 0V to VDD. The capacitive coupling caused by thegate-to-channel capacitance of the transistor T1 raises the node N1 to2×VDD−Vth with the rising of the clock signal C1. During the rising ofthe clock signal C1, the gate-to-source voltage of the transistor T1 iskept at VDD−Vth, which means the transistor T1 has a low impedance.Therefore, the output terminal OUT is charged substantially at the sametime with the rising of the clock signal C1. In other words, the outputsignal from the output terminal OUT rises following the rising of theclock signal C1, allowing the gate line GL to be activated or selected.Since the gate-to-source voltage of the transistor T1 at this timesatisfies the conditions for the transistor T1 to operate in anon-saturation region, a voltage drop by the threshold voltage (Vth)does not occur in the transistor T1, and the output terminal OUTtransitions to the same level (VDD) as the clock signal C1.

The node N1 is kept at 2×VDD−Vth until time t₅ at which the clock signalC1 returns to 0V (where a leakage current is disregarded). Therefore,the transistor T1 has a low impedance while the clock signal C1 drops inlevel, and the output terminal OUT drops in level with the falling ofthe clock signal C1. At this time, the node N1 drops from 2×VDD−Vth byVDD to VDD−Vth. Therefore, the transistor T1 is held on, and the gateline GL is brought into the inactive state having a low impedance.

At time t₆ and thereafter, the above-described operation is repeated.The gate-line driving circuit 30 needs to be operated such that gatelines GL are sequentially activated one by one in a cycle of one frameperiod. Therefore, an input signal is input to each unit shift registerSR once in one frame period. Although the operation when an input signalis input to the input terminal IN has been described above, the clocksignals C1 and C2 are continuously input in a certain cycle to the firstand second clock terminals A and B, respectively, during a period inwhich no input signal is input. Thus, the node N2 is charged repeatedlyeach time the transistor T5 is turned on in response to the clock signalC3 to keep VDD−Vth, so that the node N1 is kept at 0V. That is, duringthe period in which no signal is input, the transistors T1 and T2 ofoutput stage are held off and on, respectively, so that thecorresponding gate line GL is kept in the inactive state having a lowimpedance.

Summarizing the above-described operation, in the unit shift registerSR, the node N2 is kept at VDD−Vth during the period in which no signalis input to the input terminal IN, whereby the output terminal OUT (gateline GL) is kept at 0V having a low impedance. When a signal is input tothe input terminal IN, the node N2 transitions to 0V while the node N1is charged to VDD−Vth. Next, when the clock signal C1 is input to thefirst clock terminal A, the node N1 rises to 2×VDD−Vth, and the outputterminal OUT transitions to VDD, allowing the gate line GL to beactivated (the node N1 may thus be referred to as a “step-up node”).Thereafter, when the clock signal C3 is input to the second clockterminal B, the node N2 is reset to VDD−Vth again, and the node N1 isalso reset to 0V, so that an original state is brought about (the nodeN2 will thus be referred to as a “reset node” as well).

Connecting in cascade a plurality of unit shift registers SR operatingas described above as shown in FIG. 2 to constitute the gate-linedriving circuit 30, the input signal input to the input terminal IN ofthe unit shift register SR1 of first stage is transferred sequentiallyto the unit shift registers SR2, SR3, . . . while being shifted insynchronization with the clock signals C1, C2 and C3 as shown in atiming chart of FIG. 5. Accordingly, the gate-line driving circuit 30 iscapable of driving the gate lines GL1, GL2, GL3, . . . sequentially in apredetermined scanning cycle.

The voltage waveforms indicated by solid lines in FIG. 4 show idealwaveforms in the case where no leakage current occurs at the node N1.When a leakage current occurs at the node N1, the voltage waveforms atthe node N1 and output terminal OUT become those indicated by dottedlines in FIG. 4. That is, the level at the node N1 drops with time aftertime t₃ at which the transistor T3 is turned off. Therefore, thetransistor T1 has a high impedance at time t₅ at which the clock signalC1 drops in level, so that the voltage drop at the output terminal OUTdoes not follow the falling of the clock signal C1. That is, the leveltransition at the output terminal OUT from VDD to 0V takes time, whichin turn increases a rise time of the output signal, i.e., a drivingsignal for the gate line GL.

As shown in the lowermost position in FIG. 4, the output terminal OUT ofthe unit shift register SR of the immediately succeeding stage isactivated at time t₆. Accordingly, a plurality of adjacent gate lines GLare selected at the same time when the rise time of the output signalincreases, resulting in failure to achieve a normal display. Asdescribed earlier, the conventional unit shift register having only onetransistor connected between the node N1 and ground GND causes thisproblem when a leakage current occurs in the transistor.

In contrast, according to the present embodiment, the two transistors T4and T7 are connected in series between the node N1 and ground GND, andthe transistor T8 connected to the power source VDM is connected to thenode N3 between the transistors T4 and T7, as shown in FIG. 3. Since thegates of the transistors T4 and T7 are connected in common to the nodeN2, the transistors T4 and T7 are switched on/off with the same timing.Since the gate of the transistor T8 is connected to the node N1, thetransistor T8 is turned on when the node N1 is at a high level (that is,when the transistors T4 and T7 are held off).

Accordingly, when the transistors T4 and T7 are turned off and the nodeN1 transitions to VDD−Vth at time t₂ shown in FIG. 4 as described above,the transistor T8 is turned on, and the node N3 transitions to VDM.Focusing attention on the transistor T4 at this time, the gate (node N2)is at 0V, the drain (node N1) is at VDD−Vth, and the source (node N3) isat VDM (>0V). Therefore, the gate of the transistor T4 is negativebiased with respect to the source (this state will hereinafter bereferred to as a “negative-biased state”). This negative-biased state ofthe transistor T4 continues until the node N1 is reset to 0V (until timet₆ shown in FIG. 4). As a result, the transistor T4 is in thenegative-biased state during the period in which the node N1 is charged(from time t₂ to time t₆).

FIG. 6 is a graph showing the relationship between the gate-to-sourcevoltage (V_(GS)) and drain-to-source current (I_(DS)) in a generalN-type transistor. In FIG. 6, the vertical axis (I_(DS)) indicates alogarithmic scale. The N-type transistor is off when V_(GS)=0 holds,however, as seen from FIG. 6, a leakage current I_(OFF1) occurs whenV_(GS)=0 holds. Particularly, the leakage current I_(OFF1) is relativelyhigh when V_(GS)=0 holds in an amorphous TFT for use in a displayapparatus, and further tends to increase by one order of magnitude ormore as compared to a usual case under the influence of a backlight forimage display.

In the case of the conventional unit shift register, as the source ofthe single transistor connected between the node N1 and ground GND is atthe ground voltage GND, the gate-to-source voltage in the off state is0V. Therefore, the conventional unit shift register arises theabove-mentioned problem resulting from the leakage current I_(OFF1)occurred in the single transistor.

In contrast, the transistor T4 of the unit shift register SR accordingto the present embodiment is negative biased (V_(GS)<0) in the offstate. Assuming that V_(GS)<0 holds in an N-type transistor, a leakagecurrent I_(OFF2) at that time becomes approximately 1/1000 times theleakage current I_(OFF1) at the time when V_(GS)=0 holds.

Accordingly, the unit shift register SR according to the presentembodiment reduces a leakage current in the transistor T4 (i.e., leakagecurrent at the node N1), allowing the node N1 to be prevented fromdropping in level while being charged. This can avoid the problem inthat the output terminal OUT cannot follow the level transition of theclock signal C1. Further, since a fall time of the output signal fromthe output terminal OUT (discharge time of gate line GL) is shorter thanin a conventional gate-line driving circuit, a timing margin in thedriving operation of the gate line GL can be increased, which achievesan improved operational reliability. Accordingly, mounting the gate-linedriving circuit formed by the unit shift registers SR on the displayapparatus can prevent a malfunction and achieve a normal display.

Although FIG. 3 shows the third power terminal s3 is connected to thepower source VDM, the third power terminal s3 may be connected to thepower source VDD similarly to the first power terminal s1. In this case,there is an advantage in reducing the number of power sources required.However, some kinds of TFTs represent I_(DS)−V_(GS) characteristics asshown by dotted lines in FIG. 6. Thus, attention should be given to thecase of setting the third power terminal s3 as high as VDD, which maydegrade the effect of reducing the leakage current in the transistor T4.

As the power source VDM connected to the third power terminal s3, apower circuit which drops the source voltage VDD and outputs the droppedvoltage as the voltage VDM may be used. FIGS. 7 and 8 each illustrate anexample of such power circuit.

FIG. 7 shows a power circuit for dividing the output voltage of thepower source VDD by a transistor group DT1 formed by n units ofdiode-connected transistors connected in series and a capacitive elementCA, for generating the voltage VDM. A connection node between thetransistor group DT1 connected to the power source VDD and thecapacitive element CA connected to the ground GND is an output terminalof voltage VDM.

Since each transistor constituting the transistor group DT1 undergoes avoltage drop by the threshold voltage Vth, the voltage VDM=VDD−n×Vth isobtained at the output terminal of VDM. The capacitive element CA servesto stabilize the level of VDM against an instantaneous load current.Since a direct current hardly flows in the power source VDM with thecircuitry shown in FIG. 3, the voltage VDM can be applied from thecircuit shown in FIG. 7 to all the unit shift registers SR constitutingthe gate-line driving circuit 30.

FIG. 8 shows another example of power circuit for dropping the outputvoltage of the power source VDD to generate the voltage VDM. Atransistor group DT2 formed by three diode-connected transistorsconnected in series and a high resistive element R1 are connected inseries between the power source VDD and ground GND. A connection nodeN10 between the high resistive element R1 and transistor group DT2 isconnected to the gate of a transistor TR1. The transistor TR1 has itsdrain connected to the power source VDD and its source connected to theground GND via the capacitive element CA. A connection node between thetransistor TR1 and capacitive element CA is an output terminal of powersource VDM.

In the circuit shown in FIG. 8, the node N10 is at approximately 3×Vth,and thus, a voltage dropped from 3×Vth by a threshold voltage of thetransistor TR1 expressed as VDM=2×Vth is output to the output terminalof power source VDM. As seen from this equation, the voltage VDM doesnot depend upon the level transition of the power source VDD. Thisachieves an effect of generating the voltage VDM with more stability.This circuit is also capable of applying the voltage VDM to all the unitshift registers SR constituting the gate-line driving circuit 30,similarly to the circuit shown in FIG. 7.

Although the present embodiment has described the configuration in whichthe same power source VDD is connected to the first and second powerterminals s1 and s2, the application of the present invention is notlimited to such configuration, but different power sources may beconnected to the power terminals s1 and s2. More specifically, any powersource that outputs such a voltage that the transistors T2, T4 and T7can be turned on may be connected to the second power terminal s2instead of the power source VDD. This also applies to preferredembodiments which will be described below.

Second Preferred Embodiment

FIG. 9 is a circuit diagram illustrating the configuration of a unitshift register SR according to a second preferred embodiment. In thefirst preferred embodiment, the transistor T8 for applying the voltageVDM to the node N3 has its gate connected to the node N1, however, thetransistor T8 has its gate connected to the output terminal OUT in thepresent embodiment. That is, the transistor T8 is turned on when theoutput terminal OUT transitions to VDD.

Accordingly, in the present embodiment, the voltage VDM is applied tothe node N3 via the transistor T8 only between time t₄ and time t₅ inthe timing chart shown in FIG. 4. The node N3 is kept at VDM betweentime t₅ and time t₆ as it is in the floating state in that period. Morespecifically, in the present embodiment, the transistor T4 is in thenegative-biased state between time t₄ and time t₆, which prevents theoccurrence of leakage current at the node N1.

A period during which the occurrence of leakage current at the node N1has to be prevented is from time t₄ at which the transistor T3 is turnedoff with the node N1 being charged to time t₅ at which the clock signalC1 input to the first clock terminal A rises, however, the leakagecurrent tends to occur particularly between time t₄ and time t₅ duringwhich the node N1 rises to 2×VDD−Vth. Accordingly, the configuration ofthe present embodiment in which the transistor T4 is negative biasedonly from time t₄ to time t₅ can achieve the effect of controlling theleakage current at the node N1 at almost the same level as in the firstpreferred embodiment.

Further, in the present embodiment, the number of transistors connectedto the node N1 is less than in the first preferred embodiment, so thatthe parasitic capacitance of the node N1 is reduced. This achieves theeffect of raising the node N1 in level by the clock signal input to thefirst clock terminal A with more efficiency.

In the present embodiment, the circuit shown in FIG. 7 or 8 may be usedas means for generating the voltage of VDM.

Third Preferred Embodiment

FIG. 10 is a circuit diagram illustrating the configuration of a unitshift register SR according to a third preferred embodiment. In thepresent embodiment, the gate of the transistor T1 and the node N1 areconnected via the transistor T9. The transistor T9 has its gateconnected to a fourth power terminal s4. In the present embodiment, thefourth power terminal s4 is connected to the power source VDD, similarlyto the first and second power terminals s1 and s2. Here, a connectionnode between the gate of the transistor T1 and the transistor T9 isdefined as a node N4.

In the unit shift register SR according to the present embodiment, thenode N4 is also charged to the VDD−Vth as well as the node N1 when aninput signal is input to the input terminal IN. Then, the clock signalC1 input to the first clock terminal A transitions from 0V to VDD, andthe node N4 rises to 2×VDD−Vth by capacitive coupling caused by thegate-to-channel capacitance of the transistor T1. However, the node N1is set at a level determined by a source-follower operation of thetransistor T9. In FIG. 10, the gate voltage of the transistor T9 is VDD,and thus, the node N1 does not transition from VDD−Vth.

More specifically, in the present embodiment, the node N4 rises to2×VDD−Vth between time t₄ and time t₅ shown in the timing chart of FIG.3, while the node N1 is kept at VDD−Vth. Therefore, the drain-to-sourcevoltage of the transistor T4 between time t₄ and time t₅ is lower thanin the first preferred embodiment, achieving the effect of furtherreducing the leakage current in the transistor T4 during that timeperiod.

In the present embodiment, the gate of the transistor T9, i.e., thefourth power terminal s4 is connected to the power source VDD, similarlyto the first and second power terminals s1 and s2 to avoid an increasein the number of power sources, however, the present invention is notlimited to such configuration. Any other power source that can set thelevel at the node N1 at a value close to the level at the node N3 (VDM)by the source-follower operation of the transistor T9 may be connectedto the fourth power terminal s4. In that case, effects similar to thosedescribed above can also be achieved.

Fourth Preferred Embodiment

FIG. 11 is a circuit diagram illustrating the configuration of a unitshift register SR according to a fourth preferred embodiment. Thepresent embodiment is achieved by combining the second and thirdpreferred embodiments. More specifically, the transistor T8 has its gateconnected to the output terminal OUT, and the transistor T9 having itsgate connected to the fourth power terminal s4 is connected between thegate of the transistor T1 and node N1. The fourth power terminal s4 isconnected to the power source VDD in the present embodiment.

In the present embodiment, four transistors are connected to the node N1as shown in FIG. 10, which arises a concern about an increase inparasitic capacitance at the node N1. According to the presentembodiment, however, the transistor T8 is not connected to the node N1by applying the second preferred embodiment, which can solve suchproblem. Further, similarly to the third preferred embodiment, the nodeN1 is kept at VDD−Vth even when the node N4 rises to 2×VDD−Vth,achieving the effect of reducing the drain-to-source voltage of thetransistor T4 at that time, which can control the leakage current.

Fifth Preferred Embodiment

FIG. 12 is a circuit diagram illustrating the configuration of a unitshift register SR according to a fifth preferred embodiment. Theconfiguration of the unit shift register SR according to this embodimentis almost the same as that of the fourth preferred embodiment (FIG. 11)except that the power source VDD is connected to the third powerterminal s3 to which the transistor T8 is connected, similarly to thefirst and fourth power terminals s1 and s4.

Since the power source VDD is connected to the fourth power terminal s4to which the gate of the transistor T9 is connected, the node N1 is keptat VDD−Vth even when the node N4 rises to 2×VDD−Vth, similarly to thefourth preferred embodiment. Further, since the power source VDD isconnected to the third power terminal s3, the node N3 transitions toVDD−Vth at that time. That is, the drain-to-source voltage of thetransistor T4 transitions to approximately 0V, causing no leakagecurrent to flow between the drain and source of the transistor T4. Thisin turn achieves the effect of preventing the gate voltage of thetransistor T1 from dropping in level.

In the present embodiment, the power source VDD is connected to both thethird and fourth power terminals s3 and s4 to avoid an increase in thenumber of power sources, however, any other power source that can setthe level at the node N1 at a value almost the same as that of the nodeN3 may be used.

Sixth Preferred Embodiment

The above description has shown the configuration in which the unitshift register SR of the gate-line driving circuit 30 is operated usingthree phase clock signals C1, C2 and C3 as shown in FIG. 2, however, twophase clock signals may be used instead. FIG. 13 shows the configurationof the gate-line driving circuit 30 in that case.

In this case, the gate-line driving circuit 30 is also formed by aplurality of unit shift registers SR connected in cascade. The clockgenerator 31 outputs two phase clock signals C11 and C12 of oppositephase to each other. Either the clock signal C11 or C12 is input to thefirst clock terminal A of each unit shift register SR such that theclock signals of opposite phase to each other are input alternately toadjacent unit shift registers SR. The second clock terminal B of eachunit shift register SR receives an output signal output from a unitshift register SR of the immediately succeeding stage.

FIG. 14 is a timing chart when the gate-line driving circuit 30 isoperated using the two phase clock signals C11 and C12. An input signalinput to the input terminal IN of the unit shift register SR of firststage is transmitted sequentially to the unit shift registers SR2, SR3,. . . while being shifted in synchronization with the clock signals C11and C12. Accordingly, the gate-line driving circuit 30 is capable ofdriving the gate lines GL1, GL2, GL3, . . . sequentially in apredetermined scanning cycle.

With the circuitry shown in FIG. 13, however, the clock signal input tothe second clock terminal B of each unit shift register SR is an outputsignal from a unit shift register SR of the immediately succeedingstage. Thus, the reset node (node N2 shown in FIG. 3) is not reset toVDD−Vth until the unit shift register SR of the immediately succeedingstage is operated at least once, so that a normal operation as shown inFIG. 14 cannot be achieved. Therefore, prior to the normal operation, adummy operation needs to be performed for transmitting a dummy inputsignal from the unit shift register SR of first stage to the unit shiftregister SR of last stage. Alternatively, a transistor for reset may beprovided between the reset node and power source VDD to perform a resetoperation for previously charging the reset node prior to the normaloperation. In that case, however, a signal line for the reset needs tobe provided additionally.

The problem of leakage current in the unit shift registers SRconstituting the gate-line driving circuit 30 configured as shown inFIG. 13 will now be described. For ease of description, each of the unitshift registers SR shown in FIG. 13 is assumed to have the circuitrydescribed in the first preferred embodiment (FIG. 3).

The voltage waveform at the node N2 of the unit shift register SR1 ofthe gate-line driving circuit 30 shown in FIG. 13 is shown in thelowermost position in FIG. 14. As described above, the clock signalinput to the second clock terminal B of each unit shift register SR isan output signal from the immediately succeeding stage. Thus, the nodeN2 shall be charged only once in one frame period. That is, the node N2is in the floating state for as long as one frame period (about 16 ms),arising the need to store electric charge accumulated in that period.Thus, when the leakage current occurs at the node N2, the level at thenode N2 having been charged cannot be held for as long as one frameperiod. In this case, the impedance of the transistor T2, that is, theoutput impedance of the gate-line driving circuit 30 in thenon-selective state of the gate line GL increases, causing a problem ofunstable display.

Accordingly, this sixth preferred embodiment will propose a unit shiftregister SR capable of controlling the leakage current occurring at thenode N2.

FIG. 15 is a circuit diagram illustrating the configuration of the unitshift register SR according to the present embodiment. In the presentembodiment, the transistor T6 and ground GND (reference voltageterminal) are connected via the transistor T10. In other words, thetransistors T6 and T10 are connected in series between the node N2 andground GND. The transistor T10 has its gate connected to the inputterminal IN, similarly to the gate of the transistor T6. A connectionnode between the transistors T6 and T10 is defined as a node N5.

Further, in the present embodiment, a transistor T11 is connectedbetween the node N5 and a fifth power terminal s5. The power source VDMis connected to the fifth power terminal s5, and the transistor T11 hasits gate connected to the node N2.

As will be apparent from the above-mentioned U.S. Pat. No. 5,222,082 andJP2002-313093, the conventional unit shift register has only onetransistor connected between the node N2 and reference voltage terminal(ground GND) (e.g., transistor 21 shown in FIG. 2 of U.S. Pat. No.5,222,082). In other words, the conventional unit shift register isconfigured such that the transistor T6 has its source directly connectedto the ground GND removing the transistors T10 and T11 from the circuitshown in FIG. 15.

In the present embodiment, as shown in FIG. 15, the two transistors T6and T10 are connected in series between the node N2 and ground GND, andthe transistor T11 connected to the power source VDM is connected to thenode N5 between the transistors T6 and T10. Since the gates of thetransistors T6 and T10 are connected in common to the input terminal IN,the transistors T6 and T10 are switched on/off with the same timing.Since the gate of the transistor T11 is connected to the node N2, thetransistor T11 is turned on when the node N2 is at a high level (thatis, when the transistors T6 and T10 are off).

Accordingly, when a clock signal (output signal from the immediatelysucceeding stage) is input to the second clock terminal B of the unitshift register SR, to cause the node N2 to transition to VDD−Vth, thetransistor T11 is turned on, and the source voltage VDM is applied tothe node N5. Since the transistors T6 and T10 are held off until aninput signal is input to the input terminal IN, the node N5 transitionsto VDM. Focusing attention on the transistor T6 at this time, the gate(input terminal IN) is at 0V, the drain (node N2) is at VDD−Vth, and thesource (node N5) is at VDM (>0V). That is, the gate of the transistor T6is negative biased. This state continues until the node N2 is reset to0V in response to the input signal input to the input terminal IN.

As described, with the unit shift register SR according to the presentembodiment, the transistor T6 is in the negative-biased state while thenode N2 is charged. In this period, a leakage current in the transistorT6 is controlled based on the same theory described with respect to thetransistor T4 in the first preferred embodiment (FIG. 6). This allowsthe level at the node N2 having been charged to be held for a long time.Therefore, the present embodiment is effective in the case where thelevel at the reset node (node N2) of the unit shift register SR needs tobe held for as long as one frame period, similarly to the gate-linedriving circuit 30 shown in FIG. 13. That is, the problem of anincreased output impedance of the gate-line driving circuit 30 in thenon-selective state of the gate line GL causes unstable display can besolved.

In the present embodiment, the circuit shown in FIG. 7 or 8 may be usedas means for generating the voltage VDM.

Further, the power source VDD may be connected to the fifth powerterminal s5, similarly to the second power terminal s2. In that case,the node N5 is also charged to VDD−Vth while the node N2 is charged toVDD−Vth. More specifically, the drain-to-source voltage of thetransistor T6 transitions to approximately 0V at that time, causing noleakage current to flow in the transistor T6. This achieves the effectof preventing the node N2, i.e., the gate of the transistor T2 fromdropping in level. There is also an advantage in that the number ofpower sources required can be reduced by changing the power source VDMto power source VDD.

The above description has been based on the gate-line driving circuit 30shown in FIG. 13, however, the circuitry shown in FIG. 2 can alsoperform a normal operation and control the leakage current at the nodeN2. This also applies to unit shift registers SR according to preferredembodiments which will be described below. With the circuitry shown inFIG. 2, however, one of the clock signals C1 to C3 is input to thesecond clock terminal B, so that the node N2 is charged in that cycle,and does not remain in the floating state for as long as one frameperiod. Therefore, the problem of the leakage current at the node N2 isless serious than in the circuit shown in FIG. 13.

Seventh Preferred Embodiment

FIG. 16 is a circuit diagram illustrating the configuration, of a unitshift register SR according to a seventh preferred embodiment. In thesixth preferred embodiment (FIG. 15), the transistor T11 has its gateconnected to the node N2, however, the gate is connected to the firstclock terminal A in the present embodiment. That is, the transistor T11is turned on when the first clock terminal A transitions to VDD.

With the circuitry shown in FIG. 15, when a little leakage currentoccurs in the transistor T6, causing the node N2 to drop in level, theimpedance of the transistor T11 increases correspondingly, causing thenode N5 to also drop in level. This arises a concern that the effects ofthe present invention may be degraded so that a leakage current in thetransistor T6 increases.

In contrast, with the circuitry shown in FIG. 16, a clock signal (eitherthe clock signal C11 or C12 shown in FIG. 13) having a cycle shorterthan one frame period is input to the gate of the transistor T11. Sincethe node N5 is charged in the cycle of this clock signal withreliability, the node N5 is kept at VDM, which can prevent the effectsof the present invention from being reduced.

The first clock terminal A of each unit shift register SR receives aclock signal of opposite phase to one input to an adjacent unit shiftregister SR, and the input terminal IN receives an output signal of aunit shift register SR of the immediately preceding stage (i.e.,adjacent unit shift register SR). Accordingly, the input terminal IN andthe gate of the transistor T11 (first clock terminal A) are notactivated at the same time. Therefore, the transistors T10 and T11 arenot turned on at the same time, and a short circuit current is preventedflowing from the power source VDM to ground GND via the transistors T10and T11.

Further, in the present embodiment, the power source VDD may beconnected to the fifth power terminal s5, similarly to the second powerterminal s2. In that case, the node N5 is also charged to VDD−Vth whilethe node N2 is charged to VDD−Vth. The drain-to-source voltage of thetransistor T6 transitions to approximately 0V, so that no leakagecurrent flows in the transistor T6. There is also an advantage inreducing the number of power sources required by changing the powersource VDM to power source VDD.

Eighth Preferred Embodiment

The sixth and seventh preferred embodiments have proposed theconfiguration for controlling the leakage current in the transistor T6to solve the problem of the leakage current at the node N2 of the unitshift register SR. In contrast, to solve the problem, the presentembodiment will propose a unit shift register SR capable of compensatingfor the level transition at the node N2 caused by a leakage current.

FIG. 17 is a circuit diagram illustrating the configuration of a unitshift register SR according to an eighth preferred embodiment. As shownin the drawing, the unit shift register SR includes a compensationcircuit formed by a transistor T13 connected between a sixth powerterminal s6 and node N2, a transistor T12 connected between the gate ofthe transistor T13 (defined as a node N6) and node N2 and a capacitiveelement CB connected between the node N6 and first clock terminal A. Thetransistor T12 has its gate connected to a seventh power terminal s7. Inthe present embodiment, the first, second, sixth and seventh powerterminals s1, s2, s6 and s7 are all connected to the power source VDD.

This compensation circuit is a circuit for applying the voltage at thesixth power terminal s6 (power source VDD) to the node N2 for chargingthe node N2. More specifically, the compensation circuit applies acurrent larger than a leakage current in the transistor T6 from thesixth power terminal s6 (power source VDD) to the node N2 via thetransistor T13, thereby compensating for the level at the node N2dropped by the leakage current.

In the normal operation, the node N2 is charged to VDD−Vth when a clocksignal (output signal from the immediately succeeding stage) is input tothe second clock terminal B. Since the transistor T12 is held on at thistime, the node N6 is also charged to VDD−Vth. When the clock signal (C11or C12) at the first clock terminal A transitions from 0V to VDD afterthe second clock terminal B transitions to 0V, the node N6 risesapproximately to 2×VDD−Vth because of capacitive coupling caused by thecapacitive element CB.

Since the transistor T12 has its drain connected to the node N6 and itssource connected to the node N2, the gate-to-source voltage of thetransistor T12 is approximately Vth (threshold voltage). Thus, thetransistor T12 has a high impedance almost in the off state, so that acurrent hardly flows in the transistor T12. Accordingly, the node N6 iskept at 2×VDD−Vth while the first clock terminal A is at VDD. Since thetransistor T13 is turned on in this period, the node N2 rises to VDD.

With the circuitry shown in FIG. 13, the second clock terminal B is at0V for about one frame period, and clock signals are repeatedly input tothe first clock terminal A in that period. Since the transistor T13 isaccordingly turned on repeatedly while the second clock terminal B is at0V to charge the node N2, the level at the node N2 is compensated forand is kept at approximately VDD even when a leakage current occurs atthe node N2. In other words, the output terminal OUT can be kept at alow impedance of 0V.

When the input terminal IN transitions to VDD upon receipt of an inputsignal, the transistor T6 is turned on to cause the node N2 totransition to 0V. Then, the gate-to-source voltage of the transistor T12transitions to VDD, so that the transistor T12 is turned on, and thenode N6 transitions to a low impedance of 0V. Thus, during a period inwhich the node N2 is set at 0V, the node N6 hardly rises in level evenwhen the first clock terminal A transitions to VDD, and the transistorT13 is held off and flows no current. More specifically, the node N2does not unnecessarily rise in level when the gate line GL is selected,which prevents the transistor T2 from being turned on. In addition, ashort circuit current is prevented from flowing from the power sourceVDD to the ground GND via the transistors T13 and T6.

As described, the unit shift register SR according to the presentembodiment includes the compensation circuit for applying, to the nodeN2, a voltage (VDD in this embodiment) that holds the transistor T2 onduring a period in which the node N2 is charged to turn on thetransistor T2, thereby charging the node N2. Thus, even when a leakagecurrent occurs in the transistor T6, the level transition at the node N2is compensated for. Accordingly, the impedance of the transistor T2 isprevented from rising in the non-selective state of the gate line GL.Therefore, mounting the gate-line driving circuit formed by the unitshift register SR on the display apparatus can prevent a malfunction andachieve a normal display.

In FIG. 17, the gate of the transistor T12, that is, the seventh powerterminal s7 is connected to the power source VDD. In this case, thegate-to-source voltage of the transistor T12 just after the nodes N2 andN6 are charged transitions to Vth, bringing the transistor T12 almostinto the off state. To completely turn off the transistor T12 at thistime, the seventh power terminal s7 may be made lower than VDD, e.g.,VDD−Vth or 2×VDD−Vth.

For instance, setting the seventh power terminal s7 at VDD−Vth, thesource (node N2) is at VDD−Vth when the node N6 rises in level.Accordingly, the gate-to-source voltage of the transistor T12transitions to 0V, so that the transistor T12 is completely turned off.

Alternatively, setting the seventh power terminal s7 at VDD−2×Vth, forexample, the gate-to-source voltage of the transistor T12 transitions to−Vth when the node N6 rises in level, causing the gate to be reversebiased with respect to the source, so that the transistor T12 iscompletely turned off. In this case, the node N6 is at VDD−3×Vth beforebeing raised in level in response to the clock signal input to the firstclock terminal A, and transitions to 2×VDD−3×Vth when being raised inlevel. In other words, the gate-to-source voltage of the transistor T13is expressed as: (2×VDD−3×Vth)−(VDD−Vth)=VDD−2×Vth. Usually, VDD>>2×Vthholds, the voltage is at a level sufficient to turn on the transistorT13.

FIGS. 18 and 19 are diagrams each illustrating an example of powercircuit connected to the seventh power terminal s7. FIG. 18 shows apower circuit for generating the voltage of VDD−Vth, in which the outputof the power source VDD is output upon being divided by adiode-connected transistor DT3 and a high resistive element R2. Thecapacitive element CA is provided for stabilizing the output voltage inlevel. Since a voltage drop by the threshold voltage Vth occurs in thetransistor DT3, the output voltage of VDD−Vth is obtained from thispower circuit.

FIG. 19 shows a power circuit for generating the voltage of VDD−2×Vth,in which the output of the power source VDD is output upon being dividedby a transistor group DT4 formed by two diode-connected transistors anda high resistive element R3. Since each of the two transistorsconstituting the transistor group DT4 undergoes a voltage drop by thethreshold voltage Vth, the output voltage of VDD−2×Vth is obtained fromthis power circuit. In FIG. 19, the capacitive element CA is alsoprovided for stabilizing the output voltage in level.

Further, combining the sixth and seventh preferred embodiments with thepresent embodiment can prevent a leakage current from occurring in thetransistor T6, and a leakage current, if occurred in the transistor T6,is compensated for. Accordingly, higher effects of solving the problemof leakage current in the transistor T6 can be obtained.

Ninth Preferred Embodiment

FIG. 20 is a circuit diagram illustrating the configuration of a unitshift register SR according to a ninth preferred embodiment. Accordingto the present embodiment, as shown in the drawing, the capacitiveelement CB shown in FIG. 17 of the eighth preferred embodiment isreplaced by a capacitive element formed by a drain-source connectedtransistor T14. Such a capacitive element configured using a MOStransistor is called a “MOS capacitive element” or “channel capacitiveelement”.

In the case of replacing the capacitive element CB shown in FIG. 17 bythe MOS capacitive element formed by the transistor T14, the transistorT14 is off when the node N6 is at 0V, so that a channel is not formedbetween its source and drain, which is equivalent to that no capacitanceis connected between the node N6 and first clock terminal A. Thus, thenode N6 is kept at 0V with reliability even when the first clockterminal A transitions from 0V to VDD when the nodes N2 and N6 are at0V. That is, the transistor T13 can be turned off with reliability whenthe gate line GL is selected, which prevents an unnecessary level riseat the node N2. In other words, the transistor T2 is prevented fromturning on with more reliability when the gate line GL is selected.

Tenth Preferred Embodiment

FIG. 21 is a circuit diagram illustrating the configuration of a unitshift register SR according to a tenth preferred embodiment. The unitshift register SR replaces the transistors T5 and T6 of the circuitshown in FIG. 3 by transistors T15 to T19. That is, a driving section ofthe unit shift register SR according to the present embodiment is formedby the transistors T3, T4, T7, T8 and T15 to T19.

The transistors T15 and T16 are connected in series between the secondpower terminal s2 (power source VDD) and reference voltage terminal(ground GND), and a connection node between these transistors T15 andT16 is the node N2. The transistor T15 is diode connected, and serves asa load. The transistor T16 has its gate connected to the node N1.

The transistors T17 and T18 are connected in series between the node N1and reference voltage terminal (ground GND), and have their gatesconnected in common to the second clock terminal B (or any otherterminal that synchronizes with a clock signal input to the second clockterminal B). A connection node between these transistors T17 and T18 isdefined as a node N7. The transistor T19 is connected between the nodeN7 and an eighth power terminal s8, and has its gate connected to thenode N1. In the present embodiment, the eighth power terminal s8 isconnected to the power source VDM.

The driving section of the unit shift register SR according to thepresent embodiment has a different circuitry from those of theabove-described preferred embodiments, but operates almost in the sameway. More specifically, the driving section according to the presentembodiment also applies the voltage of the reference voltage terminal(ground GND) to the node N1 and the voltage of the second power terminals2 (power source VDD) to the node N2 in synchronization with a clocksignal input to the second clock terminal B, while applying the voltageof the first power terminal s1 (power source VDD) to the node N1 and thevoltage of the reference voltage terminal (ground GND) to the node N2 inresponse to an input signal input to the input terminal IN. Theoperation will be described below.

FIG. 22 is a timing chart showing an operation of the unit shiftregister SR according to the tenth preferred embodiment. Similarly tothe description with reference to FIG. 4, the following description willbe made assuming that the clock signal C1 is input to the first clockterminal A of the unit shift register SR and the clock signal C3 isinput to the second clock terminal B.

As shown in FIG. 22, when the clock signal C3 (at the second clockterminal B) transitions from 0V to VDD at time t₀, the transistors T17and T18 are turned on, causing the node N1 to drop in level. Then, thetransistor T16 is turned off, causing the node N2 to transition toVDD−Vth. The transistors T4 and T7 are accordingly turned on, causingthe node N1 to transition to 0V. At this time, the nodes N3 and N7 bothtransition to 0V along with the node N1. As a result, the transistor T1is turned off, and the transistor T2 is turned on, so that the outputterminal OUT transitions to 0V, bringing the gate line GL into aninactive state (non-selective state) having a low impedance.

Next, when the clock signal C3 returns to 0V at time t₁, the transistorsT17 and T18 are turned off, while the transistors T4 and T6 are held onand the transistor T16 is held off. Thus, the node N1 is kept at 0V, andthe node N2 is kept at VDD−Vth.

At time t₂, a signal is input to the input terminal IN, causing theinput terminal IN to transition to VDD. Then, the transistor T3 isturned on, causing the node N1 to rise in level. Then, the transistorT16 is turned on, causing the node N2 to transition to 0V, so that thetransistors T2, T4 and T7 are turned off. Accordingly, the node N1transitions to VDD−Vth.

In the present embodiment, the transistors T8 and T19 are turned on atthis time, causing the voltage of the power source VDM to be applied toeach of the nodes N3 and N7, so that the level at the nodes N3 and N7transition to VDM. In other words, the transistors T4 and T7 are bothbrought into the reverse biased state.

When the input terminal IN returns to 0V at time t₃, the transistor T3is turned off, however, the node N1 is brought into the floating statesince the transistors T4, T7, T17 and T18 are also off. At this time, aleakage current hardly occurs at the node N1 since the transistors T4and T7 are both reverse biased. Accordingly, the node N1 is kept atVDD−Vth with reliability.

Next, when the clock signal C1 input to the first clock terminal Atransitions from 0V to VDD at time t₄, capacitive coupling caused by thegate-to-channel capacitance of the transistor T1 allows the gate of thetransistor T1 to rise in level following the rising of the clock signalC1, causing the node N1 to transition to 2×VDD−Vth. The output terminalOUT transitions to VDD following the rising of the clock signal C1, sothat the gate line GL is activated.

At time t₅, the clock signal C1 transitions to 0V. Since a leakagecurrent hardly occurs at the node N1, the node N1 is kept at 2×VDD−Vthuntil this point of time, and the output terminal OUT drops to 0Vfollowing the falling of the clock signal C1.

At time t₆ and thereafter, the above-described operation is repeated.The gate-line driving circuit 30 operates such that gate lines GL aresequentially activated one by one in a cycle of one frame period.Therefore, an input signal is input to each unit shift register SR onlyonce in one frame period. The clock signals C1 and C3 are also input tothe first and second clock terminals A and B, respectively, during aperiod in which no signal is input (i.e., in the non-selective state ofthe gate line GL). In this period, the transistors T4 and T7 are heldon, and the transistor T16 is held off, so that the node N1 is kept at0V, and the node N2 is kept at VDD−Vth. Therefore, in the non-selectivestate of the gate line GL, the transistor T1 is held off, and thetransistor T2 is held on.

For instance, the unit shift registers SR described in the first tofifth preferred embodiments cause a problem in that the node N2 cannotbe kept at VDD−Vth when a leakage current occurs at the node N2 sincethe node N2 is brought into the floating state when the second clockterminal B transitions to 0V during the period in which no signal isinput. As described earlier, particularly when connecting a plurality ofunit shift registers SR as shown in FIG. 13, the period during which thenode N2 is in the floating state is one frame period, making the problemmore serious. Accordingly, in the present invention, the sixth to ninthpreferred embodiments each have proposed the unit shift register SR thatcan solve the problem.

In contrast, in the unit shift register SR according to the presentembodiment, once the node N1 is set at 0V and the node N2 at VDD−Vth,the transistors T4 and T7 are held on and the transistor T16 is held offuntil the input terminal IN next transitions to VDD. Thus, the node N2is kept at VDD−Vth without being brought into the floating state. Inother words, the transistors T3, T4, T7, T5 and T16 serve as a flip-flopcircuit, and the node N1 is latched at 0V and the node N2 is latched atVDD−Vth. Therefore, the present embodiment brings an advantage in thatthe above-mentioned problem resulting from the leakage current at thenode N2 does not arise. However, power consumption is higher than in thefirst to ninth preferred embodiments since a short circuit current flowsfrom the power source VDD to ground GND via the transistors T15 and T16during a period in which an input signal is input to the input terminalIN and the node N2 is set at 0V (from time t₂ to time t₆ shown in FIG.22).

As described above, the driving section of the unit shift register SRaccording to the present embodiment is configured such that thetransistors T8 and T19 are turned on and the voltage of the power sourceVDD is applied to both the nodes N3 and N7 during a period in which thenode N1 transitions to VDD−Vth (i.e., a period in which the transistorsT4, T7, T17 and T18 are held off: in the present embodiment, from timet₂ to time t₆ shown in FIG. 22). That is, the transistors T4 and T17interposed between the node N1 and ground GND are reverse biased in thatperiod, allowing the leakage current at the node N1 to be reduced.

Therefore, according to the present embodiment, the node N1 is preventedfrom dropping in level while being charged. This can avoid the problemin that the output terminal OUT cannot follow the level transition ofthe clock signal C1, similarly to the first preferred embodiment.Further, the fall time (discharge time of the gate line GL) of theoutput signal from the output terminal OUT is shorter than in theconventional gate-line driving circuit, which produces an effect ofproviding a greater timing margin in the driving operation of the gateline GL.

Although FIG. 21 shows that the power source VDM is connected to thethird and eighth power terminals s3 and s8, the power source VDD may beconnected instead, similarly to the first power terminal s1. In thiscase, there is an advantage in reducing the number of power sourcesrequired. However, some kinds of TFTs represent I_(DS)−V_(GS)characteristics as shown by dotted lines in FIG. 6. Thus, attentionshould be given to the case of setting the third and eighth powerterminals s3 and s8 as high as VDD, which may degrade the effect ofreducing leakage currents in the transistors T4 and T17.

Further, the circuit (formed by the third power terminal s3 andtransistor T8) for applying the source voltage VDM to the node N3 andthe circuit (formed by the eighth power terminal s8 and transistor T19)for applying the source voltage VDM to the node N7 are providedseparately in the present embodiment. However, connecting the nodes N3and N7 in common as shown in FIG. 23, only one circuit is required forapplying the source voltage VDM to the nodes N3 and N7, achieving areduction in circuit scale. In FIG. 23, the transistor T8 applies thevoltage at the third power terminal s3 to both the nodes N3 and N7. Theabove-described operation is possible since the transistors T4, T7, T17and T18 are all held off during a period in which the transistor T8 isheld on (i.e., a period in which the node N1 is at VDD−Vth).

Eleventh Preferred Embodiment

FIG. 24 is a circuit diagram illustrating the configuration of a unitshift register SR according to an eleventh preferred embodiment. In thetenth preferred embodiment, the transistors T8 and T19 for applying thevoltage VDM to the nodes N3 and N7 have their gates connected to thenode N1, however, the gates are connected to the output terminal OUT inthe present embodiment. More specifically, the transistors T8 and T19are turned on when the output terminal OUT transitions to VDD.

Therefore, in the present embodiment, the voltage VDM is applied to thenodes N3 and N7 only from time t₄ to time t₅ in the timing chart of FIG.22. However, the nodes N3 and N7 are brought into the floating statefrom time t₅ to time t₆, and are kept at VDM in that period. That is,the transistors T4 and T17 are negative biased from time t₄ to time t₆in the present embodiment, preventing a leakage current from occurringat the node N1.

Accordingly, the present embodiment achieves the effect obtained by thesecond preferred embodiment. That is, the effect of controlling aleakage current at the node N1 at almost the same level as in the firstpreferred embodiment. Further, the number of transistors connected tothe node N1 is less than in the tenth preferred embodiment. Thus, theparasitic capacitance at the node N1 is reduced, achieving the effect ofraising the node N1 in level by the clock signal input to the firstclock terminal A with more efficiency.

Furthermore, although not shown, the nodes N3 and N7 may be connected incommon in the present embodiment. In this case, only one circuit isrequired for applying the source voltage VDM to the nodes N3 and N7,achieving a reduction in circuit scale.

Twelfth Preferred Embodiment

FIG. 25 is a circuit diagram illustrating the configuration of a unitshift register SR according to a twelfth preferred embodiment. Thepresent embodiment applies the technique described in the thirdpreferred embodiment to the unit shift register SR according to thetenth preferred embodiment. More specifically, the unit shift registerSR according to the present embodiment is configured such that the gateof the transistor T1 (node N4) and node N1 of the circuit shown in FIG.21 are connected via the transistor T9. The fourth power terminal s4 towhich the gate of the transistor T9 is connected is connected to thepower source VDD, similarly to the first and second power terminals s1and s2.

In this unit shift register SR, the node N1 is set at the leveldetermined by the source-follower operation of the transistor T9 duringa period in which the gate of the transistor T1 (node N4) rises to2×VDD−Vth (from time t₄ to time t₅ in the timing chart of FIG. 22). InFIG. 25, the gate voltage of the transistor T9 is at VDD, so that thenode N1 is kept at VDD−Vth. Therefore, the drain-to-source voltage ofthe transistor T4 (the voltage between the nodes N1 and N3) and thedrain-to-source voltage of the transistor T17 (the voltage between thenodes N1 and N7) are lower than in the tenth preferred embodiment,achieving an effect of further reducing a leakage current in thetransistor T4 in that period.

In the present embodiment, the gate of the transistor T9, i.e., thefourth power terminal s4 is connected to the power source VDD, similarlyto the first and second power terminals s1 and s2. However, any otherpower source that can set the node N1 at a value close to the level atthe nodes N3 and N7 (VDM) by the source-follower operation of thetransistor T9 may be used. In that case, similar effects as describedabove can also be achieved.

Thirteenth Preferred Embodiment

FIG. 26 is a circuit diagram illustrating the configuration of a unitshift register SR according to a thirteenth preferred embodiment. Thepresent embodiment is achieved by combining the eleventh and twelfthpreferred embodiments. More specifically, the transistors T8 and T19have their gates connected to the output terminal OUT, and thetransistor T9 having its gate connected to the fourth power terminal s4is interposed between the gate of the transistor T1 and node N1. In thepresent embodiment, the fourth power terminal s4 is connected to thepower source VDD.

In the above twelfth preferred embodiment, seven transistors areconnected to the node N1 as shown in FIG. 25, causing a concern aboutincreasing the parasitic capacitance at the node N1. In the presentembodiment, however, the transistors T8 and T19 are not connected to thenode N1 by applying the eleventh preferred embodiment, so that theproblem is solved. Further, similarly to the twelfth preferredembodiment, the node N1 is kept at VDD−Vth even when the node N4 risesto 2×VDD−Vth. Thus, the drain-to-source voltage of each of thetransistors T4 and T19 at that time is reduced, achieving the effect ofcontrolling leakage currents in the transistors T4 and T19.

Fourteenth Preferred Embodiment

FIG. 27 is a circuit diagram illustrating the configuration of a unitshift register SR according to a fourteenth preferred embodiment. Theunit shift register SR is configured such that the power source VDD isconnected to the third and eighth power terminals s3 and s8, similarlyto the first and fourth power terminals s1 and s4, by applying the fifthpreferred embodiment to the circuit according to the thirteenthpreferred embodiment (FIG. 26).

Since the power source VDD is connected to the fourth power terminal s4to which the gate of the transistor T9 is connected, the node N1 is keptat VDD−Vth even when the node N4 rises to 2×VDD−Vth, similarly to thethirteenth preferred embodiment. Further, since the power source VDD isconnected to the third and eighth power terminals s3 and s8, the node N3also transitions to VDD−Vth at that time. In other words, thedrain-to-source voltage of each of the transistors T4 and T17transitions to approximately 0V, so that no leakage current flowsbetween the drain and source of each of the transistors T4 and T17. As aresult, the effect of preventing the node N4, i.e., the gate of thetransistor T1 from dropping in level is obtained.

In the present embodiment, the power source VDD is connected to thethird, fourth and eighth power terminals s3, s4 and s8 to avoid anincrease in the number of power sources, however, any other power sourcemay be used that can set the nodes N1, N4 and N7 at almost the samevalue when the node N1 rises in level.

Fifteenth Preferred Embodiment

For instance, the unit shift register SR according to the firstpreferred embodiment is configured as shown in FIG. 3 such that thetransistor T4 is negative biased using the power source VDM connected tothe third power terminal s3 and transistors T7 and T8, therebycontrolling the leakage current at the node N1. The present embodimentwill propose a unit shift register SR capable of bringing the transistorT4 into the negative-biased state without using these components.

FIG. 28 is a circuit diagram illustrating the configuration of a unitshift register SR according to a fifteenth preferred embodiment. In thepresent embodiment, the node N3, i.e., the source of the transistor T4is connected to the output terminal OUT. As described above, the thirdpower terminal s3 (power source VDM) and transistors T7 and T8 shown inFIG. 3 are not required in this circuit.

As seen from FIG. 3, the transistors T2 and T7 are switched on/off withthe same timing as their sources are both connected to the ground GNDand their gates are connected to the node N2. Therefore, a normaloperation can be performed similarly to the circuit shown in FIG. 3 byconnecting the transistor T2 (in place of transistor T7) between thenode N3 and ground GND, as shown in FIG. 28.

However, in the unit shift register SR shown in FIG. 28, the node N3 isconnected to the output terminal OUT, so that the node N3 alsotransitions to VDD when the output terminal OUT transitions to VDD. Inother words, the unit shift register according to the present embodimentoperates such that the node N3 transitions to VDD during a period fromtime t₄ to time t₅ in the timing chart of FIG. 4. Accordingly, thetransistor T4 is negative biased in that period, which can control theleakage current at the node N1.

Referring to the timing chart shown in FIG. 4, the period during whichthe leakage current at the node N1 should be prevented is from time t₄at which the transistor T3 is turned off with the node N1 being chargedto time t₅ at which the clock signal C1 input to the first clockterminal A falls, however, the leakage current tends to occurparticularly from time t₄ to time t₅ during which the node N1 rises to2×VDD−Vth. Therefore, the effect of controlling the leakage current atthe node N1 at almost the same level as in the first preferredembodiment is obtained even with the circuitry of the present embodimentin which the transistor T4 is negative biased only from time t₄ to timet₅.

Further, in the present embodiment, the number of required transistorsand power sources is less than in the first preferred embodiment,achieving a reduction in circuit scale. The number of transistorsconnected to the node N1 is also less than in the first preferredembodiment, so that the parasitic capacitance at the node N1 is reduced.This can achieve an effect of raising the node N1 in level by the clocksignal input to the first clock terminal A with more efficiency.

Sixteenth Preferred Embodiment

FIG. 29 is a circuit diagram illustrating the configuration of a unitshift register SR according to a sixteenth preferred embodiment. Thepresent embodiment applies the technique described in the fifteenthpreferred embodiment to the unit shift register SR according to thetenth preferred embodiment.

In the present embodiment, the node N3 (the source of the transistor T4)and the node N7 (the connection node between the transistors T17 andT18) are both connected to the output terminal OUT. The third and eighthpower terminals s3 and s8 (power source VDM) and transistors T7, T8 andT19 shown in FIG. 21 are not required.

Connecting the transistor T2 (in place of the transistor T7) between thenode N3 and ground GND arises no problem in the operation, as describedin the fifteenth preferred embodiment.

On the other hand, paying attention to the behaviors of the transistorsT2, T17 and T18 in the normal operation of the unit shift register SRshown in FIG. 22, the transistor T2 is also turned on when thetransistors T17 and T18 are turned on, and the transistors T17 and T18are turned off when the transistor T2 is turned off to cause the outputterminal OUT to transition to VDD. Therefore, connecting the node N7 tothe output terminal OUT arises no problem in the operation.

In the normal operation, the node N1 needs to be set 0V in response tothe clock signal input to the second clock terminal B, so that thetransistor T18 cannot be omitted. This is because, although beingconnected between the node N1 and ground GND, the transistor T2 is firstturned on when the node N1 drops in level to turn on the transistor T16and to raise the node N2 in level, as described in the tenth preferredembodiment, so that the node N1 cannot be discharged substantially viathe transistor T2.

In the unit shift register SR shown in FIG. 29, the nodes N3 and N7 areconnected to the output terminal OUT, and thus transition to VDD whenthe output terminal OUT transitions to VDD. More specifically, the unitshift register SR according to the present embodiment operates such thatthe nodes N3 and N7 transition to VDD between time t₄ and time t₅ in thetiming chart shown in FIG. 4. Therefore, the transistors T4 and T17 arenegative biased in that period, which controls the leakage current atthe node N1.

Referring to the timing chart shown in FIG. 22, the period during whichthe leakage current at the node N1 should be prevented is from time t₄at which the transistor T3 is turned off with the node N1 being chargedto time t₅ at which the clock signal C1 input to the first clockterminal A falls, however, the leakage current tends to occurparticularly from time t₄ to time t₅ during which the node N1 rises to2×VDD−Vth. This achieves an effect of controlling the leakage current atthe node N1 at almost the same level as in the fifteenth preferredembodiment even with the circuitry of the present embodiment in whichthe transistors T4 and T7 are negative biased only from time t₄ to timet₅.

Further, in the present embodiment, the number of required transistorsand power sources is less than in the tenth preferred embodiment,achieving a reduction in circuit scale. The number of transistorsconnected to the node N1 is also less than in the tenth preferredembodiment, so that the parasitic capacitance at the node N1 is reduced.This achieves an effect of raising the node N1 in level by the clocksignal input to the first clock terminal A with more efficiency.

Seventeenth Preferred Embodiment

In a display apparatus, there is a possibility that noise from the dataline DL due to coupling caused by the parasitic capacitance between thegate line GL and data line DL, for example, may be added to the outputterminal OUT of the unit shift register SR in the non-selective state ofthe gate line GL.

For instance, in the unit shift register SR according to the fifteenthpreferred embodiment (FIG. 28), the node N2 is at VDD−Vth in thenon-selective state of the gate line GL, so that the transistor T4 ison. When noise from the gate line GL is added to the output terminal OUTat that time, the noise is transmitted to the node N1 via the transistorT4. If the transistor T1 is thereby turned on, a corresponding gate lineGL is activated though not being selected, which may cause a malfunctionof failing to achieve a normal display.

FIG. 30 is a circuit diagram illustrating the configuration of a unitshift register SR according to a seventeenth preferred embodiment. Asshown, the node N3 is not connected to the output terminal OUT in theunit shift register SR.

A transistors T21 is connected between the node N3 and first clockterminal A, and a transistor T22 is connected between the node N3 andground GND (reference voltage terminal). More specifically, the pair oftransistors T21 and T22 and that of the transistors T1 and T2 areconnected in parallel to each other. The transistor T21 has its gateconnected to the node N1, similarly to the transistor T1, and thetransistor T22 has its gate connected to the node N2, similarly to thetransistor T2. Except this configuration, the unit shift register SR ofthe present embodiment is configured similarly to that shown in FIG. 28.

The transistors T21 and T22 carry out the same operation as thetransistors T1 and T2, respectively, so that the node N3 and outputterminal OUT transition in level just in the same way. As a result, theunit shift register SR shown in FIG. 30 carries out the same operationas the unit shift register SR according to the fifteenth preferredembodiment. More specifically, in the present embodiment, the transistorT4 is negative biased from time t₄ to time t₅ in the timing chart shownin FIG. 4, which controls the leakage current at the node N1.

In the present embodiment, however, the output terminal OUT and node N3are separated, different from the fifteenth preferred embodiment.Therefore, noise, if added from the gate line GL to the output terminalOUT, is prevented from being transmitted to the node N1, which can avoidthe above-mentioned malfunction.

Eighteenth Preferred Embodiment

The present embodiment applies the technique described in theseventeenth preferred embodiment to the unit shift register SR accordingto the sixteenth preferred embodiment (FIG. 29).

FIG. 31 is a circuit diagram illustrating the configuration of a unitshift register SR according to an eighteenth preferred embodiment. Asshown, in the unit shift register SR, the node N3 and output terminalOUT are not connected.

Similarly to the seventeenth preferred embodiment, the transistor T21having its gate connected to the node N1 is connected between the nodeN3 and first clock terminal A, and the transistor T22 having its gateconnected to the node N2 is connected between the node N3 and ground GND(reference voltage terminal). Except this configuration, the unit shiftregister SR of the present embodiment is configured similarly to thatshown in FIG. 29.

The transistors T21 and T22 carry out the same operation as thetransistors T1 and T2, respectively, so that the node N3 and outputterminal OUT transition in level just in the same way. As a result, theunit shift register SR shown in FIG. 31 carries out the same operationas the unit shift register SR according to the sixteenth preferredembodiment. More specifically, in the present embodiment, the transistorT4 is negative biased from time t₄ to time t₅ in the timing chart shownin FIG. 4, which controls the leakage current at the node N1.

In the present embodiment, however, the output terminal OUT and node N3are separated, different from the sixteenth preferred embodiment.Therefore, the above-mentioned malfunction caused by noise added fromthe gate line GL to the output terminal OUT can be avoided.

While the invention has been shown and described in detail, theforegoing description is in all aspects illustrative and notrestrictive. It is therefore understood that numerous modifications andvariations can be devised without departing from the scope of theinvention.

1. A shift register comprising: an input terminal and an outputterminals; first and second clock terminals to which first and secondclock signals are input, respectively, said first and second clocksignals being shifted in phase from each other; first, second and thirdvoltage terminals to which first, second and third voltages are applied,respectively; a first transistor connected between said output terminaland said first clock terminals; a second transistor connected betweensaid output terminal and said first voltage terminal; a first node towhich a control electrode of said first transistor is connected; asecond node to which a control electrode of said second transistor isconnected; and a driving section for applying said first voltage to saidfirst node and a voltage corresponding to said third voltage to saidsecond node in synchronization with said second clock signal, whileapplying a voltage corresponding to said second voltage to said firstnode and said first voltage to said second node in response to an inputsignal input to said input terminal, wherein said driving sectionincludes third and fourth transistors connected in series between saidsecond node and said first voltage terminal, each of said third andfourth transistors having a control electrode connected to said inputterminal, and said driving section is configured such that apredetermined voltage other than said first voltage is applied to athird node which is a connection node between said third and fourthtransistors in a period during which said third and fourth transistorsare in a cutoff state.
 2. The shift register according to claim 1,wherein said driving section further includes: a fourth voltage terminalto which a predetermined fourth voltage is applied; and a fifthtransistor connected between said fourth voltage terminal and said thirdnode.
 3. The shift register according to claim 2, wherein said fifthtransistor has a control electrode connected to said second node.
 4. Ashift register comprising a plurality of shift registers connected incascade, each being defined in claim
 1. 5. An image display apparatususing said shift register defined in claim 1 as a scanning-line drivingcircuit.
 6. The shift register according to claim 1, wherein said secondvoltage and said third voltage are at the same level.
 7. The shiftregister according to claim 2, wherein said third voltage and saidfourth voltage are at the same level.
 8. The shift register according toclaim 2, wherein said fifth transistor has a control electrode connectedto said first clock terminal.